Measuring 2 nV/√Hz noise and 120 dB supply rejection in linear regulators; the Quest for Quiet, part 3

Measuring 2 nV/√Hz noise and 120 dB supply rejection in linear regulators; the Quest for Quiet, part 3

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By eeNews Europe

Part 1 of this article is here, and part 2 here.

Measuring regulator output noise

Once the amplifier is checked and calibrated, actual noise measurements are made. Accurately measuring linear regulator output noise and obtaining faithful results requires careful attention to DUT shielding, component choice, layout, and cable management.

Figure 9. Noise measurement bench setup. Shielded box houses noise amplifier, low output impedance of linear regulator removes necessity for shielding, but magnetic fields can still affect output


Figure 9 shows the configuration used for testing a linear regulator, highlighting the construction and shielding used to avoid magnetic fields from intruding on the measurement. Only one instrument is connected at any given time to preclude ground loops from corrupting the measurement.

Battery power is chosen to supply the linear regulator for the same reason as powering the amplifier; the goal is to measure the noise of the linear regulator, not characterise supply rejection. The regulator does not need to be shielded, as the low output impedance of the regulator makes it much less susceptible to low frequency magnetic fields. Connections from the regulator output to the noise amplifier need to be short barrel connectors since long flexible cables will introduce errors due to triboelectric [4] effects.

The amplifier output is fed directly into an oscilloscope to measure peak-to-peak noise. As shown in Figure 10, the peak-to-peak noise of the LT3042 is 4 µVP-P. A spectrum analyser plot of the same regulator (shown in Figure 11) shows the noise for various amounts of SET pin capacitance. The RMS noise from 10 Hz to 100 kHz as a function of SET pin capacitance is shown in Figure 12.

Figure 10. LT3042 noise in 10 Hz to 100 kHz bandwidth. RMS noise measures 0.8 μVRMS

Figure 11. Noise spectral density plot shows effect of increasing SET pin capacitance on LT3042

Figure 12. Increasing SET pin capacitance decreases RMS noise in 10 Hz to 100 kHz bandwidth



Measuring RMS noise requires one to be more fastidious in selecting instrumentation. All RMS voltmeters are not created equal, please review Appendix C, “Understanding and Selecting RMS Voltmeters”, in [Linear Technology’s] AN83, “Performance Verification of Low Noise, Low Dropout Regulators,” for information on types of RMS voltmeters and their performance. This appendix lists many different RMS voltmeters and highlights how some have significant errors, leading to measurements that are more optimistic than reality.

Just as important as noise; measuring supply rejection

Supply rejection for a linear regulator is just as critical as output voltage noise. With poor supply rejection, even the lowest noise regulator will pass signal through to the output, and this can swamp the noise from the regulator. Switching regulators are often employed as pre-regulators to provide an optimal combination of efficiency, noise, transient response, and output impedance.

Most state-of-the-art switching regulators operate at frequencies from 100 kHz to 4 MHz. Even with the lowest ESR capacitors, the pulsed nature of energy transfer that defines switching regulators creates output voltage ripple at the switch frequency. These signals cause problems in noise-sensitive video, communications, and other types of circuitry. This has been touched on in Linear Technology Application Note 101, “Minimizing Switching Regulator Residue in Linear Regulator Outputs,” published in July of 2005.

Recently released linear regulators are promising supply rejection in the range of 80 dB and above. The LT3042 approaches 120 dB of supply rejection at certain frequencies. For testing this, the input must be kept at a low enough amplitude to ensure the small signal response of the regulator is tested as opposed to large signal response, although enough signal must be generated to have a measurable signal at the output. Additionally, the input DC level with the superimposed AC signal must not drive the regulator into dropout or other unwanted operation regions.

Driving the DUT

When testing regulator rejection, the first thing that must be done is to supply a signal to be rejected. This is more complex than just connecting a frequency generator to the device; the AC signal must ride on top of a DC offset and be capable of providing the current required under load.

Figure 13. Driver board sums AC and DC voltages to provide several Amps at frequencies to 10 MHz. A high resolution version of this figure is here.


The circuit used for this purpose was developed by Jim Williams and is shown in Figure 13. In this circuit, a DC reference voltage is generated by A2 and summed with an AC signal on the inverting input of A1. The output of A1 drives Darlington connected transistors that are paralleled with ballast resistors to drive up to 5A output current.

One major caveat needs to be noted when connecting this circuit to the DUT: input capacitance for the regulator should not be used. The first reason is that the circuit is not optimised for driving capacitive loads and may oscillate. Second, there is no ability for this circuit to sink current; a load must be present to discharge input capacitors, especially as frequency increases. Driving a 50 mVP-P sinusoidal signal at 10 MHz across a 1µF capacitor requires over 3A of charge and discharge current to prevent signal distortion. If making measurements at light output currents (under 100 mA), use a preload to ensure signal fidelity presented to the regulator.


Thinking ahead

When a device presents very high supply rejection, careful attention must be paid to the details of the instrumentation. If the regulator provides 100 dB of supply rejection, a 50 mVP-P input signal is reduced to 0.5 µVP-P at the output. Increasing the input signal amplitude is possible, but at some point a transition from small signal response to large signal will occur.

For a regulator with high supply rejection, the low amplitude of the output signal is comparable to, or smaller than, the amplitude of the noise of the device. This suggests that we should amplify the signal much the same that we did for the noise to be able to make accurate measurements. Even with this, the output signal is often hidden by the noise. Fortunately, modern oscilloscopes provide averaging capabilities that allow one to extract the signal from the noise; the average value of random noise is zero. The input signal supplies the trigger needed.

Whether or not the signal is amplified, other possible issues arise when measuring supply rejection. The input and output signals must be measured simultaneously; one needs both input and output amplitudes to know the rejection of the device. A block diagram of the measurement setup is reviewed in Figure 14.

Figure 14. Block diagram of supply rejection measurement shows ground loops. Switching to differential to single-ended amplifier resolves ground loops. A high resolution version of this figure is here.

Worth noting in the block diagram is that ground loops exist that can corrupt the measurement. The first is the ground loop formed through the common ground of the two oscilloscope channels. This loop passes through the signal amplifier and any signal in the ground loop corrupts the supply rejection measurement, giving results that do not reflect actual performance. The solution to this is to switch the signal amplifier from a single-ended circuit to fully differential. In doing so, both loops are broken and measurement fidelity returns. The second loop (not shown on Figure 14) comes through the AC line ground to the first oscilloscope channel. This loop shows minimal contribution to errors as all signals are large in comparison.

Simple amplifier for differential inputs

A simple amplifier is shown in Figure 15. This amplifier uses a fully differential gain stage on the input with a gain of 40 dB followed by a differential to single-ended converter to give another 20 dB of gain. Each input has a 200 Hz highpass filter to block DC. The LTC6409 was chosen for its high gain-bandwidth product of 10 GHz. The second stage is provided by an LT1818 configured as a differential to single-ended converter with a gain of 20 dB.

Figure 15. Simple differential to single-ended amplifier provides 60 dB of gain


The input referred noise of this combination of amplifiers runs approximately 1.4 nV/√Hz, which means that we should expect less than 2.2 µVP-P of noise. At the same time, we expect 4 µVP-P of noise from the regulator itself. Compared to the 0.5 µVP-P of signal we are expecting at the output of the regulator, this noise completely swamps the signal we are trying to measure. Again, the saving grace is the random nature of noise giving an average value of zero: using a modern oscilloscope with memory, averaging reveals the signal hidden within the noise.

Improved differential amplifier

Measurements from extremely high performing linear regulators get even trickier. With only 60 dB of gain on the output signal, a 0.5 µVP-P signal becomes 0.5 mVP-P. This small amplitude is approaching the measurement thresholds of many high end oscilloscopes with 1X probes. Increasing the input amplitude to the linear regulator tenfold gives added headroom, but if regulator supply rejection increases another 20 dB, then the problem once again rears its head.

Figure 16. Improved amplifier provides differential inputs with 80 dB of gain. A high resolution version of this figure is here.


Figure 16 shows the implementation of a higher performance amplifier. It is based on both the noise amplifier of Figure 2 and the previous differential to single-ended amplifier in Figure 15. Now, the LT1818 used for each stage is replaced with LT1994 differential amplifiers that feed back to the differential transistor pairs, still formed by the THAT300 transistor arrays. A second stage of differential gain comes from another LT1994 before being converted to single-ended measurement through the first LT6232. Successive stages for highpass and Butterworth filters follow those in Figure 2. Calibration and verification of circuit response is the same as the low noise amplifier.


Figure 17. Setup for measuring supply rejection. Driver board and DUT on bottom left, amplifier board on bottom right. Power supplies and signal source not shown


Figure 18. Supply rejection plot of LT3042 shows >70 dB performance to frequencies approaching 4 MHz


The setup to measure supply rejection is shown in Figure 17. Measured supply rejection of the LT3042 regulator is shown in Figure 18. It is worth noting that the supply rejection of the regulator approaches 120 dB at 100 Hz. Verification of this measurement on an oscilloscope necessitates the 80 dB of gain from the improved amplifier.

The final part of the article will look at alternative measurements, measurement pitfalls, and present appendices on magnetic shielding materials, and on dealing with high frequency switching artifacts.

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