Measuring 2 nV/√Hz noise and 120 dB supply rejection in linear regulators; the Quest for Quiet, part 4

Measuring 2 nV/√Hz noise and 120 dB supply rejection in linear regulators; the Quest for Quiet, part 4

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Part 1 of this article is here, part 2 here, and part 3 here.

Other methods for measurement

Other methods and equipment are available to make supply rejection measurements. A lock-in amplifier uses the reference signal to provide synchronous detection at the desired frequency to help measure small signal levels. A network analyser also provides an oscillator to sweep across frequency while providing the bandpass function to measure both input and output amplitudes and calculate the rejection of the circuit. These methods provide valid results, but one still needs to be fastidious with circuit connections and verify results. Checking both input and output signals on an oscilloscope is essential; signal amplitudes and wave shapes will indicate if the regulator under test is being driven into dropout or if small signal response has given way to large signal behaviour.


Similar to measuring noise, there are pitfalls that can lead one astray when measuring supply rejection. Careful attention to circuit grounding using star grounds is important. Some effects that are seen while measuring supply rejection actually seem counter-intuitive.

Up until now, a solid design would always include some capacitance at the input of the linear regulator to keep the supply impedance as low as possible across frequency. With high enough supply rejection from a device, this can actually increase the amount of ripple seen at the output.

Figure 19. Using the LT3042 to post-regulate the LT8614 Silent Switcher

Consider a circuit as shown in Figure 19 where the LT3042 post-regulates the LT8614 Silent Switcher regulator. The LT8614 delivers approximately 20 mVP-P of ripple at its 500 kHz switching frequency to the input of the LT3042 through a few cm of copper board traces. With only the 22 µF output capacitor of the LT8614, output ripple of the linear regulator is only a few µVP-P. When a 4.7 µF capacitor is added at the input of the LT3042, output ripple increases to approximately 75 µVP-P, as shown in Figure 20. It should be noted that bandwidth was limited to 20 MHz for these photos as the goal was to show ripple at the switching frequency, not high frequency edge spikes.



Figure 20. The LT3042 post-regulating the LT8614 Silent Switcher (a) without any capacitor at the LT3042 input, (b) with a 4.7 μF capacitor at LT3042 input. Both photos are bandwidth limited to ignore high frequency spikes

How is it that adding input capacitance reduces the supply rejection of the regulator? The answer lies not with the performance of the LT3042, but in the board layout. The LT3042 provides exceptional capability at electrically rejecting signals from the input supply. Up until now, the ability to reject these signals was the limiting factor. Now, magnetic fields become the culprit.

To better understand this, the schematic in Figure 21 highlights an AC current path of the DC-DC converter with a solid green line. If capacitance is present at the input of the LT3042, AC current also flows in the broken green path. The input of the LT3042 presents a high impedance at the frequency of concern, so no AC current flows into the LT3042.

Figure 21. Schematic highlights AC current loop of DC-DC converter together with paths susceptible to magnetic coupling

The AC current flow creates a magnetic field that will produce current in other nearby loops, the same way one winding of a transformer couples to other windings. Two loops of concern are shown in blue and red on Figure 21. The blue loop, formed by CSET and RSET, generates ripple at the input to the error amplifier. With the unity-gain architecture of the LT3042, this ripple is transferred through to the output. The red loop formed by the output capacitor and the impedance looking back into the regulator (and nearby load components as well) generates ripple directly on the regulator output.

Counter intuitively, removing capacitance from the input of the LT3042 reduces output ripple. Given that this is not electrical feedthrough of signal, but is instead magnetic coupling, one must think about distance, shielding, and loop orientation when designing boards. Field strength is related to distance and loop area, so minimising loop area (by not using an input capacitor) and maximising distance (by only using the DC-DC converter output capacitance) limits the current impressed on sensitive loops.

This reveals that the decision early on to not use capacitors on the output of the signal driver board or input of the regulator was prudent. With a capacitor right at the inputs of the regulator, a loop is added creating magnetic fields that couple into the output and give erroneous results. The regulator supply rejection would appear to be much worse than it actually is.

One other issue that comes to light when using switching regulators is not just removing the switch frequency ripple, but the spikes associated with the switch edges. Switch edges on some circuits are transitioning in just a few nanoseconds, translating to frequency content in the 100s of MHz. These frequencies cannot be easily eliminated with a simple linear regulator. Parasitic effects such as trace capacitance and magnetic coupling make reduction of these spikes difficult. Please refer to Linear Technology Application Note 101, “Minimizing Switching Regulator Residue in Linear Regulator Outputs,” and Appendix B, Controlling High Frequency Switch Spikes, for further information.


The exacting performance offered by linear regulators such as the LT3042 provides exceptionally quiet supply rails for sensitive systems. Verification of DC performance from such a device is usually not a tricky proposition. Critical parameters such as noise and supply rejection are not as easy to measure at such high performance levels. Careful attention must be paid to the smallest of details in measurement circuits, connections, board layout, and equipment. What were once small errors that could be ignored (compared to the signal being measured) are now first-order error terms. The high PSRR performance delivered shows that signals are not being transmitted through the device itself, but through magnetic coupling. Every detail must be scrutinized to ensure fidelity of measurements and provide solid results that are trustworthy.

Todd Owen and Amit Patel are Senior IC Design Engineers with Linear Technology Corporation.


1. Morrison, Ralph, “Grounding and Shielding Techniques in Instrumentation,” Wiley-Interscience, 1986.

2. Ott, Henry W., “Noise Reduction Techniques in Electronic Systems,” Wiley-Interscience, 1976.

3. THAT 300 Series Data Sheet, THAT Corporation.

4. Williams, Jim, “775 Nanovolt Noise Measurement for A Low Noise Voltage Reference,” Linear Technology Corporation, Application Note 124, July 2009.

5. Williams, Jim and Owen, Todd, “Performance Verification of Low Noise, Low Dropout Regulators,” Linear Technology Corporation, Application Note 83, March 2000.

6. Patel, Amit, “Industry’s First 0.8µVRMS Noise LDO Has 79dB Power Supply Rejection Ratio at 1MHz,” Linear Technology Journal of Analog Innovation, April 2015, pp. 1-7.

7. Williams, Jim, “Practical Circuitry for Measurement and Control Problems,” “Symmetrical White Gaussian Noise” Appendix B, Linear Technology Corporation, Application Note 61, August 1994, pp. 38-39.

8. Williams, Jim, “Minimizing Switching Regulator Residue in Linear Regulator Outputs,” Linear Technology Corporation, Application Note 101, July 2005.

9. Metglas 2705M Technical Bulletin, Metglas, Inc.

10. Metglas 2714A Technical Bulletin, Metglas, Inc.

11. ZG-2 Brochure, Magnetic Shield Corporation.

12. Mu-2 MuMETAL Brochure, Magnetic Shield Corporation.

13. Sandler, Steven and Hymowitz, Charles. “Capacitor Values: Don’t Believe the Label,” Power Electronics Technology, May 2007, pp. 22-27.

14. Sikula, J., J. Hlavka, J. Pavelka, V. Sedlakova, L. Grmela, M. Tacano, and S. Hashiguchi. “Low Frequency Noise of Tantalum Capacitors.” Active and Passive Electronic Components 25.2 (2002): 161-67. Web.

15. Kueck, Christian, “Power Supply Layout and EMI,” Linear Technology Corporation, Application Note 139, October 2012.

Appendix A

Materials for magnetic field shielding

Aluminium and copper enclosures are commonly used for RF shielding to keep unwanted signals out of sensitive circuitry. These enclosures will not work with magnetic coupling, especially at the low frequencies seen in bench testing. A thin walled steel can (even enclosing a copper or aluminium box) will not provide enough attenuation of magnetic fields. What is needed is a material with very high magnetic permeability—the lines of magnetic flux must be diverted around the circuit instead of being allowed to go through it. Common techniques use multiple layers of materials of shielding separated by an air gap. Each subsequent layer helps attenuate the field strength as well as providing some distance between layers for further reduction. The problem lies in trying to look at magnetic permeability of materials and determine the thickness and configuration needed to shield the circuit adequately from fields. A simple steel enclosure would require extremely thick walls (estimates are at least 1/2-in. – 1.25 cm – wall thickness) in order to provide adequate shielding. This would require a custom steel box welded together, one that would most likely never move once placed on a bench due to the weight involved. Depending on the alloy, steel has a relative magnetic permeability between 400 and 2000, while copper and aluminium are very close to 1.

Materials with extremely high magnetic permeability became a requirement for shielding. High-nickel alloys such as Mu Metal offer a relative magnetic permeability of 20,000 ranging to 50,000 when properly annealed after forming. Testing chambers are available from Mu Metal provider Magnetic Shield Corporation that comprise three nested cylinders; one places sensitive circuitry inside to avoid magnetic coupling issues. These chambers are expensive, but are a simple solution to provide magnetic shielding. If you do not have an issue with creating your own enclosure, sheets of Mu Metal are available for purchase for cutting and forming to custom shapes. Thicker sheets are recommended for best shielding to avoid saturation of the material and magnetic field incursion. Be aware that the relative magnetic permeability will drop during forming and the material must be properly annealed afterwards in a hydrogen-rich atmosphere.

Another possible material is an amorphous metal alloy called Metglas. Metglas arrives in two-inch (5-cm) wide ribbons that are 0.8 mils (0.02 mm/20 microns) thick. It is not available in large sheets to cut and form into enclosures as is Mu Metal. It is used by wrapping the ribbon around the board with the overlap being enough to prevent unwanted fields from getting through; magnetic fields tend to penetrate to a depth of only twice the diameter of any hole in the material. Metglas offers much higher levels of relative magnetic permeability than Mu Metal depending on the alloy. Metglas 2705M was tried as it gives an as-cast relative magnetic permeability of 290,000 without needing special annealing techniques in a hydrogen-rich atmosphere. If an annealing chamber is available, Metglas 2714A offers a relative magnetic permeability that reaches 1,000,000. The issue found with the use of Metglas for shielding lies in the 0.8 mil thickness; the thin ribbon is easier to saturate and multiple layers are required for effective shielding.

Appendix B

Controlling high frequency switch spikes

Some of the latest switch-mode power supply circuits have much faster switch transition times compared to previous generations. The faster transitions have benefits that be seen in improved efficiency and smaller external components among others. The difficulty with these faster edges comes in minimising the associated spikes that make their way onto and into sensitive circuitry.

Switch transition times have sped up, with harmonic content going from the tens of MHz to frequencies now approaching 1 GHz. Designers in the RF realm will appreciate the difficulty in manipulating these signals. At such frequencies, component parasitics dominate, traces become transmission lines, and antennas now exist that transmit and receive energy around the board. There are two ways that high frequency content is injected into the output of a linear regulator: conducted and magnetically coupled. Treating the problem in the RF realm provides the best results in taming the issue.

Conducted signals make their way through the regulator and onto the output. The linear regulator itself cannot actively reject any of these signals; the unity-gain bandwidth of the regulator is usually a maximum of 1 MHz. The regulator has a parasitic capacitance from input to output that allows high frequency content to propagate through. Parasitic inductance and resistance degrade the effectiveness of output capacitors at these frequencies.

The best way to control conducted spikes is through the use of ferrite beads on the input of the regulator. In the frequency range where the regulator is active, the ferrite bead provides a low loss path. As frequency rises above this range, the bead increases in impedance, limiting the high frequency energy passed through.

Controlling the paths for radiated signals to make their way to the output requires attention to detail the same as with conduction paths. Ferrite beads will not prevent radiated energy from coupling onto the output of the regulator. Shielding offers a good way to help minimise high frequency radiated content from coupling onto the linear regulator output. Additionally, providing separation from components that see high frequency signals reduces field strength.

Thinking about both paths for high frequency signals to make their way onto the output will provide best results in creating a quiet supply for sensitive circuitry. A combination of ferrite beads, shielding, and distance will offer optimum overall rejection. The linear regulator and its output capacitor should be shielded with an appropriate material and placed physically away from the ‘hot’ loop of the switching supply. The input capacitor of the linear regulator should be located close to the switching supply instead of next to the linear regulator. Two to three inches (5 – 8 cm) of distance is enough to reduce the fields associated with the AC currents through the capacitor. Lastly, a ferrite bead should be placed between the switching supply and the regulator. The ferrite bead is placed between the input capacitor and the regulator without deleterious effects. Figure B1 shows a schematic of how to connect the circuit, while Figure B2 highlights construction details.

Figure B1. Use of ferrite bead, shielding, and physical distance combine to provide optimum high

frequency spike reduction

Figure B2. Construction of board from Figure B

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